1. Field of the Invention
This invention relates to communications, and, more particularly, to a system and method for a bias network of a power amplifier.
2. Description of Related Art
An ideal power amplifier amplifies an input signal with no waveshape alteration. The ideal power amplifier is therefore characterized as having a transfer function (input signal vs. output signal) which is linear with no transfer function discontinuities. In practice, however, a power amplifier has a transfer function with nonlinear and “linear” regions. Whether the power amplifier is operating in a linear or nonlinear region depends in part on the amplitude of the input signal. For the power amplifier to achieve as near to linear operation as possible, the power amplifier is designed to operate within its linear region given the range of possible input signal amplitudes. If the input signal has an amplitude which causes the power amplifier to operate outside the linear region, the power amplifier introduces nonlinear components or distortion to the signal. When the input signal possesses peak amplitudes which cause the amplifier to compress, to saturate (no appreciable increase in output amplitude with an increase in input amplitude) or to shut-off (no appreciable decrease in output amplitude with a decrease in input amplitude), the output signal is clipped or distorted in a nonlinear fashion. Generally, an amplifier is characterized as having a clipping threshold, and input signals having amplitudes beyond the clipping threshold are clipped at the amplifier output. In addition to distorting the signal, the clipping or nonlinear distortion of the input signal, generates spectral regrowth or adjacent channel power (ACP) that can interfere with an adjacent frequency.
In wireless communications systems, high power amplification of signals for transmission are commonly encountered with very large peak to average power ratios (PAR). For example, in a time division multiple access (TDMA) system, such as Global System for Mobile Communications (GSM) or North American TDMA, when multiple carrier signals are combined for amplification with a power amplifier, the resulting PAR is about 9–10 dB for a large number of carriers. In a code division multiple access (CDMA) system, a single loaded 1.25 Mhz wide carrier can typically have a PAR of 11.3 dB. For orthogonal frequency division multiplexing (OFDM), multicarrier signals can have a PAR of up to 20 dB. These signals have to be amplified fairly linearly to avoid generating ACP.
Due to the potential for high peak powers in wireless communications signals, CDMA, TDMA and frequency division multiple access (FDMA) base stations typically use radio frequency (RF) amplifiers operating in class AB mode and biased with a high current to be able to handle those peak powers. The efficiency of these amplifiers is typically less than 10%. This low efficiency leads to higher power consumption, lower overall reliability and higher operating temperatures. For low power levels, the amplifier will operate in class A operation, which has poor power added efficiency (PAE). But for the big signal peaks the amplifier will operate in class AB, which will have better PAE. A class A type amplifier is biased as a current source that conducts over a full 360° of input signal, while a class AB amplifier has a conduction angle, α, which is between 180° and 360°. The conduction angle, α, is an indication on the proportion of the RF cycle of which conduction of the transistor(s) is(are) conducting. A class A amplifier has a conduction angle of α=360° and is often operated as a small signal amplifier. In practice, it is preferred to have a class AB operation where the conduction of the transistors in the amplifier has a conduction angle less than 360°. This reduction in conduction angle translates to a bigger swing in DC supply current passed through the bias circuits, from the large energy-storing capacitors to the drain terminal of the amplifier device, for the large signal peaks (which can be as high as 12 dB higher than the rms signal power).
Because the efficiency of the amplifier is inversely related to its linearity, various linearization methods are used to enable the use of more cost-effective and more power efficient amplifiers while maintaining an acceptable level of linearity. Feed-forward correction is routinely deployed in modem amplifiers to improve the linearity of the main amplifier with various input patterns. The essence of the feed-forward correction is to isolate the distortion generated by the main amplifier on a feed forward path. The distortion is provided to a correction amplifier on the feed forward path which amplifies the distortion. The distortion on the feed forward path is combined with the distortion on the main signal path to cancel the distortion on the main signal path.
Predistortion techniques work by applying the inverse transfer characteristics of the amplifier to the signal before the signal is applied to the amplifier. The simplified mathematical explanation of the predistortion concept can be expressed as:s(t)·h(s)·y(s)=s(t),where s(t) is the time domain signal, h(s) is the amplifier transfer characteristics curve and y(s) is the inverse of the amplifier characteristics curve y(s)=1/h(s). The predistortion is important when the amplifier operates in the more non-linear region class AB, whereas the amplifier's FET's are biased for class A operation for small signals. When the amplifier operates in class AB, which is for large peaks close to maximum power of the amplifier, the current (at the drain terminal) will be pulsing with frequency components up to the envelope bandwidth. The drain current Ids will be “switched” on-off at a frequency equal to the signal envelope. The maximum current-frequency will be limited by the envelope frequency or bandwidth of the input signal at the gate of the FET device. In practice, the predistortion added to the signal widens the bandwidth or envelope of the input signal to the amplifier, for example creating a bandwidth expansion factor of 2.5 to 3. The wideband linearity of the predistortion amplifier depends on the ability to model the non-linearity of the amplifier. This is hampered if dynamic peak clipping occurs. Dynamic peak clipping occurs when a voltage drop across the DC bias network occurs when the amplifier draws large currents at the signal peaks and hence maximum available power is reduced below the expected available power.
Today, the gate and drain bias networks for the amplifier use a ladder-type of bias circuit. A ladder-type of bias circuit has a series-parallel-series parallel structure with any combination of resistors, capacitors and inductors. FIG. 1 shows a ladder tupe bias circuit 20 with series inductors 22a and 22b, parallel capacitors 24a–c and a parallel inductor 26. Traditionally, the bias networks of an amplifier have been designed to be as narrow as possible to filter out any signals beyond a few hundred kilohertz or a few megahertz to prevent those signals, including the RF or signal envelope (baseband) signals, from going into the DC bias networks. Many times a series device in the DC bias networks helps to narrow the bias circuit filter. However, a series device, such as an inductor or ferrite bead, on the DC bias network can have a significant impedance at baseband frequency (or the envelope frequency or bandwidth of the input signal). As such, there will be a voltage drop across that element as a function of the current. This voltage drop can harm the amplifier in two ways:
1) the series voltage drop will causes the amplifier to have less supply bias voltage Vdd, hence the saturated output power level P1 dB will decrease which will cause increased harmonic and intermodulation distortion (IMD); and
2) the series voltage drop will become a voltage signal (compared to the current pulsing at envelope frequencies) resulting in the drain voltage being modulated and mixing with the RF input signal, hence creating higher IMDs.
Additionally, in multiple stage amplifier configurations having a driver stage and an amplification stage, the DC bias networks for the driver stage amplifier and the amplification stage amplifier needs to be isolated from frequencies within the input signal baseband or envelope frequencies. FIG. 2a shows a multiple stage amplifier 28 having first and second stages 29a and b with corresponding DC bias networks 30aand 30b for each stage. If a series voltage drop is to occur at either or both bias networks 30a or b, leakage of the low frequency signals, for example up to a few hundred kilohertz or a few Megahertz, within the envelope frequency can be coupled in between the stages, leading to higher IMDs. In a multiple amplifier configuration using a common power bus, these signals could adversely effect other amplifiers. FIG. 2b shows the bias networks 30a and 30b used with a parallel stage amplifier 31.
Narrow bias network designs have worked fine with the use of narrowband signals, such as AMPS, TDMA/IS-136, and GSM signals, where the modulation envelope frequency is low. As the wideband radio concept has caught on, the traditional feed-forward amplifier, with its extended and somewhat costly circuitry to handle the cancellation of the inter-modulation distortion (IMD), has not had any problems with wider bandwidth signals, as long as the correction loop amplifier circuitry could handle the bandwidth.
However, as the bandwidth of signals gets wider and the use of bandwidth widening predistortion techniques are used, impedence variations of the DC bias network across the input signal envelope frequency band lead to voltage drops within the input signal envelope frequency band in current bias network structures. Such voltage drops lead to dynamic peak clipping where amplifier draws large currents at the signal peaks and hence maximum available power is reduced below the expected available power, thereby degrading the wideband linearity of the amplifier.
FIG. 3 shows the real and imaginary impedances of a bias network from the drain terminal of an LDMOS amplifier with the short-circuiting of the power input to the bias network. As shown, the real impedance curve 32 and the imaginary impedance curve 33 both show a rise and fall in the impedance of the bias network across the input signal envelope frequency band. For example, the real impedance reaches a peak 34 at about 600 KHz while the imaginary impedance crosses from a positive to negative value, a classical example of a tuned-resonant circuit. This resonance within the input signal envelope frequency cause voltage variations across the bias network at such resonant frequencies. Such voltage variations degrades the amplifier performance and reduces the effectiveness of predistortion linearization techniques because the power supplied by the bias network to the amplifier changes with the changing input signal, thereby changing the characteristics of the amplifier being predistorted and leading to nonlinear components being introduced into the amplifier output.